In a typical communications system, a transmitter encodes data to be transmitted and this encoded data is used to modulate a carrier signal which is transmitted over a communications channel to a receiver. The receiver demodulates the received carrier signal to obtain the encoded data and thereafter decodes the encoded data to obtain the original data sent by the transmitter. The type of modulation utilized in the communications system indicates the parameter of the carrier signal that is varied to encode the data onto the carrier signal, such as amplitude, frequency, or phase modulation and combinations thereof, as will be understood by those skilled in the art. One such modulation technique is known as frequency division multiplexing which utilize, instead of a single carrier signal, a number of subcarrier signals at different frequencies that are simultaneously communicated over the communications channel. With frequency division multiplexing a portion of the data being transmitted is modulated onto each of the subcarrier signals.
A frequency division multiplexing modulation technique known as orthogonal frequency division multiplexing (OFDM) has become increasingly popular in recent years for use in wireless communications systems. This is true because the intrinsic characteristics of OFDM allow it to handle the most common types of distortion found in the wireless environment, such as fading of the received signal resulting from the transmitted signal arriving at the receiver over multiple propagation paths due to reflections and objects present between the transmitter and receiver. As a result, many wireless local area networks (WLANs) utilize OFDM in the form of the IEEE 802.11 family of standards that are more commonly referred to as “wireless fidelity” or “Wi-Fi.”
In a wireless OFDM communications system, an OFDM signal is formed through the summation of a number of orthogonal subcarriers, each subcarrier having a different frequency and itself being modulated through a particular modulation technique such as quadrature amplitude modulation (QAM). The subcarriers are orthogonal in that the dot product of each subcarrier with any of the other subcarriers is equal to zero, as will be understood by those skilled in the art. This orthogonal property of each subcarrier results in a frequency spectrum for each subcarrier that has a peak frequency component FC positioned between peak frequencies f of adjacent subcarriers and that has frequency nulls positioned at the locations of these adjacent peak frequencies as shown in FIG. 1. The resulting frequency spectrums for these orthogonal subcarriers enable each of these subcarriers to be reliably demodulated from the OFDM signal, with each subcarrier thereafter being demodulated through the appropriate technique, such as QAM, to retrieve the underlying data, as will be understood by those skilled in the art.
The OFDM signal formed by the summation of these orthogonal subcarriers is typically utilized to modulate a radio frequency (RF) carrier signal which, in turn, is the actual signal propagated over the wireless channel. More specifically, a local oscillator in the transmitter typically generates the RF carrier signal and a modulation circuit modulates this carrier signal with the OFDM signal. In the receiver, a local oscillator generates a signal that is applied to a mixer along with the received modulated RF carrier signal to thereby remove the carrier signal portion and provide the original OFDM signal in the receiver.
Although OFDM is well suited for use in wireless communications systems as previously mentioned, there are nonetheless types of distortion that must be compensated for with OFDM just as with any modulation technique. One such type of distortion is frequency-offset distortion that results from a difference between the frequency of an oscillator or RF signal generated by the local oscillator in the OFDM transmitter and the frequency of the RF signal generated by the local oscillator in the OFDM receiver. Such a frequency offset results in the OFDM signal having a frequency offset relative the original OFDM signal in the transmitter, and this type of distortion can adversely affect the performance of the system.
To overcome the problem of frequency offset, the receiver typically includes a carrier tracking circuit that adjusts the frequency of the local oscillator in the receiver to match that of the local oscillator in the transmitter as closely as possible. FIG. 2 is a functional block diagram illustrating a conventional carrier tracking circuit 200 contained in an OFDM receiver for adjusting the frequency of a signal generated by a numerically controlled oscillator (NCO) 202 in the receiver. Although not shown in FIG. 2, the modulated RF signal received from the transmitter (not shown) is first demodulated or “down converted” using a local oscillator signal from a local oscillator (not shown) contained in the receiver. As previously mentioned, ideally the frequency of the local oscillator signal in the receiver is the same as that of the local oscillator in the transmitter that was modulated by the OFDM signal.
Once the RF signal has been down converted to provide the OFDM signal in the OFDM receiver, this OFDM signal is sampled to generate a plurality of samples r(n). The NCO 202 then adjusts the phases of each of these samples to compensate for a frequency offset between the local oscillator in the receiver and the local oscillator in the transmitter, as will now be explained in more detail. Note that OFDM signal in the receiver generated after down conversion is not identical to the OFDM signal in transmitter if there is a frequency offset between the local oscillator in the transmitter and the local oscillator in the receiver. Such a frequency offset results in a linearly cumulative phase error for each sample r(n) of the OFDM signal, as will be understood by those skilled in the art. As result, if the first sample r(n) has a phase error θerr then the second sample will have a phase offset 2θerr, and so on for each sample. Thus, each sample may be viewed as having a phase offset of n times θerr where n indicates the sample number.
The NCO 202 receives a phase adjustment value PHADJ from a loop filter 204, with the PHADJ value having a value corresponding to the incremental amount by which the phase of each of the samples r(n) is to be adjusted. A summation circuit 206 adds the PHADJ value with a phase increment value PHINC output from a delay circuit 208 to generate an accumulated phase value PHAC that is also input to the delay circuit. The delay circuit 208 outputs the PHAC value on its input as the PHINC value on its output a sample delay time τs later. A second summation circuit 210 subtracts the PHINC value from a phase of the sample r(n) to generate a phase-corrected sample {dot over (r)}(n) that is supplied to a fast Fourier transform (FFT) circuit 212. The sample delay time τs corresponds to time between which the samples r(n) are applied to the summation circuit 210.
Once the FFT 212 receives a required number N of the phase-corrected input samples {dot over (r)}(n), the FFT executes a fast Fourier transform algorithm to calculate a set of complex frequency components R(1)-R(N). The fast Fourier algorithm is a discrete Fourier transform algorithm that greatly reduces the number of computations required to compute the discrete Fourier transform of the input signal corresponding to the N samples {dot over (r)}(n). Each of the frequency components R(1)-R(N) is a complex number giving a magnitude and phase for a corresponding sinusoidal component of the input signal in a given frequency range or “bin” of the FFT 212. Recall that according to Fourier transform theory, a signal may be represented as a summation of sinusoids of different frequencies and varying magnitudes and phases. The frequency components R(1)-R(N) generated by the FFT 212 define the magnitudes and phases of N sinusoids, each at a corresponding frequency, which, when summed together yield the input signal corresponding to the phase-corrected samples {dot over (r)}(n). In the just described operation of the FFT 212, a single OFDM symbol that encodes corresponding data is represented by the N phase-corrected input samples {dot over (r)}(n).
The carrier tracking circuit 200 further includes a phase error estimator 214 that generates an estimated phase error value EPHE from the frequency components R(1)-R(N) output by the FFT 212. The estimator 214 typically utilizes frequency components R(1)-R(N) having known phase values to calculate the EPHE value. For example, in an OFDM system that complies with the IEEE 802.11 standard, N=64 so that the FFT 212 includes 64 bins and outputs the frequency components R(1)-R(64) in the respective bins. Of these 64 bins, four bins contain what are typically referred to as “pilot tones” having known amplitude and phase values. The phase error estimator 214 determines the estimated phase error value EPHE from phase values of the frequency components R(1)-R(64) corresponding to the pilot tones. The loop filter 204 generates the phase adjustment value PHADJ having a value that is a fraction of the EPHE value from the estimator 214. The loop filter 204 feeds back only a fraction of the EPHE value in the form of the PHADJ signal to improve the stability of the circuit 200 since the NCO 202 utilizes the PHADJ value to ultimately adjust the phase of each of the samples r(n), as will be appreciated by those skilled in the art.
In operation, assume that the delay circuit 208 initially outputs a zero value for the PHINC value and that the samples r(n) corresponding to a first OFDM symbol are sequentially applied to the summation circuit 210. Also assume the PHADJ value from the loop filter 204 has an initial default value PHADJ-DF and that the first sample r(1) of the first OFDM symbol is applied to the summation circuit 210 within the delay time τs of the delay circuit 208 after the PHADJ-DF value is initially input to the delay circuit. At this point the delay circuit 208 outputs a zero for the PHINC value when the r(1) sample is applied to the summation circuit 210 since the PHADJ-DF value is not yet output from the delay circuit. The summation circuit 210 therefore subtracts zero from the phase of the first sample r(1) and in this way does not adjust the phase value of this sample. Thus, the summation circuit 210 outputs the sample r(n) with no phase correction as the phase-corrected sample {dot over (r)}(1) that is supplied to the FFT 212.
After the delay time τs, the delay circuit 208 outputs the PHADJ-DF value as the phase increment value PHINC that is applied to the summation circuit 210. This occurs at the same time the second sample r(2) of the current OFDM symbol is applied to the summation circuit 210, causing the circuit 210 to subtract the PHINC value from the phase of this sample. The summation circuit 210 at this point outputs the phase-corrected sample {dot over (r)}(2) to the FFT 212, with this phase-corrected sample corresponding to the sample r(2) having the PHINC value subtracted from it phase. Note that while the summation circuit 210 is adjusting the phase of the sample r(2), the output of the delay circuit 208 is also fed back and applied to the summation circuit 206. As a result, the summation circuit 206 sums the PHINC value, which is equal to PHADJ-DF at this point, and the PHADJ-DF value from the loop filter 204 and outputs a value of 2×PHADJ-DF as the accumulated phase value PHAC that is supplied to the delay circuit 208.
After the delay time τs, the delay circuit 208 outputs the value 2×PHADJ-DF as the PHINC value that is supplied to the summation circuit 210, which occurs at the same time the sample r(3) is being applied to the summation circuit. The summation circuit 210 therefore subtracts the value 2×PHADJ-DF from the phase of the sample r(3) to thereby supply the phase-corrected sample {dot over (r)}(3) to the FFT 212. The NCO 202 continues operating in this manner for all samples r(n) in the first OFDM symbol, and in this way linearly increases the phase offset PHINC that the circuit 210 subtracts from the phase of the sequential samples r(1), r(2), r(3), and so on to r(N). Accordingly, the phase offset of zero is subtracted from the sample r(1), the phase offset subtracted from sample r(2) is PHADJ-DF, phase offset subtracted from sample r(3) is 2×PHADJ-DF, phase offset subtracted from sample r(4) is 3×PHADJ-DF, and so on until the phase offset of (N−1)×PHADJ-DF is subtracted from the final sample r(N) of the OFDM symbol.
Once the summation circuit 210 supplies the phase-corrected sample {dot over (r)}(N) to the FFT 212, the FFT calculates the corresponding frequency components R(1)-R(N) for this first OFDM symbol from the phase-corrected samples {dot over (r)}(1)-{dot over (r)}(N). The FFT 210 outputs these components R(1)-R(N) from the carrier tracking circuit 200 to other circuitry (not shown) in the OFDM receiver for processing. At the same time, the phase error estimator 214 generates a new estimated phase error value EPHE from the ones of these components R(1)-R(N) corresponding to the pilot tones in the OFDM symbol. As previously mentioned, in an IEEE 802.11 system each OFDM symbol includes four pilot tones having known phase values and the estimator 214 compares the actual to the expected phase values for these pilot tones to generate four phase error components, and then takes the average of these four phase error components to generate the EPHE value.
In response to this new EPHE value from the estimator 214, the loop filter 204 takes a fraction of this value to generate the new PHADJ value that is applied to the summation circuit 206. The summation circuit 206 sums this new PHADJ value with the current PHINC value to generate a new PHAC value, which is thereafter output from the delay circuit 208 the delay time τs when the first sample r(1) of the next OFDM symbol is supplied to the summation circuit 210. This is true if the delay through the FFT 212 is relatively short such that this updated value for the EPHE may be fed back to the NCO 202 in time to apply this updated value to the samples r(n) of the next OFDM symbol. In reality, however, the delay through the FFT 212 may be several symbols long such that, eventually, the EPHE calculated for a given OFDM symbol, say symbol X, is used to correct the phases of samples r(n) in an OFDM symbol X+Y where Y is the number of symbol delays through the FFT.
Because the delay time through the FFT 212 can be relatively long, the carrier tracking circuit 200 is always using the estimated phase error EPHE from a prior OFDM symbol to correct the phase errors of samples r(n) for a subsequent OFDM symbol. In fact the FFT 212 may have a pipelined architecture in which several OFDM symbols are applied to the FFT before the frequency components R(1)-R(64) corresponding to a first OFDM symbol are output by the FFT, as alluded to above. In this situation, the phase error EPHE from an initial OFDM symbol is used to correct the phase errors of samples r(n) for an OFDM symbol that is sampled several symbols after that initial symbol. This may result in inaccurate phase adjustments of the samples r(n) in the subsequent OFDM symbol due to the intervening linearly cumulative phase error of the samples.
It should also be noted that in the circuit 200 the phase error estimator 214 is used to estimate an average phase error EPHE that is then used to adjust the phases of the samples r(n) of a subsequent OFDM symbol. Thus, if this average phase error EPHE is considered to be the phase error at the middle one of the samples, which is r(32) in the IEEE 802.11 example currently being used, then an additional 32 samples will have occurred since this estimated phase error value. As a result, the phase adjustment of the NCO 202 will adjust the phases of the samples r(n) of the next OFDM symbol based upon the PHADJ value which was generated from this average phase error value EPHE. This means that the NCO 202 will adjust the phase of the first sample r(n) of the next OFDM symbol by the PHADJ value corresponding to the middle sample r(32) of the prior OFDM symbol, even though thirty two more samples have occurred between this middle sample and the first sample of the next symbol. Recall, the phase shift error from one sample r(n) to another is linearly cumulative due to the frequency offset, and thus the use of this average phase error value PHADJ does not take into account the linearly accumulated phase error between sample location where the error was estimated and the current sample. Note that while the carrier tracking circuit 200 includes the FFT 212 and is discussed as being part of an OFDM receiver, the concepts discussed above apply generally to other types of systems as well. More specifically, the concepts apply to any carrier tracking or other similar circuit where an element (e.g., the FFT 212 in FIG. 2) introduces a delay between the NCO 202 and an output of the element from which a phase error is estimated.
There is a need for a carrier tracking circuit and method that more quickly and more accurately correct for phase errors of a received OFDM or other type of signal due to frequency offset distortion.